In radio-frequency engineering, “RF power amplifiers” are normally used on the output side in the transmission path. In the case of third-generation mobile radios, for example, which are designed for UMTS (Universal Mobile Telecommunications Standard), “W-CDMA” (Wide-Band Code Division Multiple Access) signals with a bandwidth of two megahertz need to be amplified over an exponential characteristic curve on a carrier-frequency level of approximately 2 Ghz. An exponential transfer characteristic of this type is typical of variable gain amplifiers (VGAs) and can be plotted as a straight line in a semilogarithmic representation. In this case, the output power in decibels is normally plotted over the control voltage of the amplifier in volts.
Adjustable radio-frequency amplifiers of this type are subject to great demands in terms of current drawn, efficiency, linearity, noise and chip area requirement. The latter demand results in problems with crosstalk, and in radio-frequency engineering, particularly in unwanted break-through of carrier frequencies, self-mixing effects etc. The current drawn normally needs to be small because such mobile radios are normally powered from batteries or storage batteries.
Power amplifiers in the radio-frequency range are normally in the form of inductively degenerated differential amplifiers which are designed using bipolar circuitry and have a connected current cascode.
A differential amplifier of this type is specified, by way of example, in the document P. R. Gray, R. G. Meyer “Analysis and Design of Analog Integrated Circuits”, John Wiley and Sons, 1993, pages 377 to 378 and 511 to 513, cf. FIG. 5–10 therein, for example. This differential amplifier can be used to advantage, particularly with respect to the high gain, the high cut-off frequencies, low inherent noise and good linearity on account of the inductive degeneration. In this case, the output of the cascode stage is normally in the form of an open collector output in integrated transmission circuits. As a load resistor, a surface acoustic wave filter is normally connected to a power amplifier stage of this type, which represents the nonreactive load thereof. This serves to suppress unwanted signal components. Any adaptive network which may be required at the output of the open collector output is normally connected externally in this case.
The inductive degeneration of the differential amplifier with inductances connected between the common emitter node of the amplifier and the emitter connections of the two differential amplifier transistors brings about the negative current feedback which is wanted, said negative current feedback giving rise to just low inherent noise in the overall system when the quality is high and allowing the transfer function to be linearized.
When amplifying radio-frequency signals, however, the inductance disadvantageously acts as an antenna or transformer, particularly in the GHz range, and thus presents an unavoidable problem as regards the crosstalk of signals through the alternating electromagnetic field on the integrated circuit onto other circuits. A further problem is regulation of the power output from the amplifier. For the gain A, it holds true that A is proportional to the product of gm,red and Rload, where gm,red represents the transconductance of the common-emitter differential amplifier transistors. Rload is the load resistance connected to the open collector output. In this case, the gain of the cascode stage has been assumed to be one. The gradient is essentially determined by the collector current and the value of the emitter degeneration in the relevant radio-frequency range, in line with the following rule:
      g          m      ,      red        ≈      1                  1                  g          m                    +              Z        L              ≈      1                            U          T                          I          C                    +              Z        L            where gm is the gradient, ZL is the complex degeneration resistance, IC is the collector current and UT is the voltage equivalent of thermal energy.
It can be seen that the power output from the amplifier can be controlled only by intervening in the latter's supply of quiescent current, which would disadvantageously shift the amplifier's operating point accordingly. With a relatively large gain range, however, this would bring about unwanted nonlinearities in the differential amplifier.
The document A. B. Grebene “Bipolar and MOS Analog Integrated Circuit Design”, John Wiley and Sons 1984, pages 444 to 448, illustrates a developed amplifier with a current shunting principle in the cascode stage. In this case, two cascode stages are connected in parallel (cf. FIG. 8.37 therein, for example) which change over the current signal from the amplifier smoothly from the output to the supply voltage. This is possible if an additional control block is used to shift the bias potentials of the cascode transistors linearly. As a result,
      I    C    =            I      S        ·          exp      ⁡              (                              U            BE                                U            T                          )            results in an exponential characteristic curve from the amplifier for regulation which is linear in a semi-logarithmic representation. The most serious drawback of power regulation of this type is provided by the constant current drawn by the power amplifier. A power-saving mode can be provided only by using a not insignificant amount of auxiliary power for the control block. The amplifier itself requires a constant power at every point of the characteristic curve, which significantly reduces the efficiency of the overall circuit when the output power is brought down.